Method and system for simultaneously broadcasting and receiving digital and analog signals

ABSTRACT

A system and method for transmitting digital information through a medium such as atmospheric free-space includes a transmitter which generates a signal based on a basis set of mutually orthogonal, spectrally-shaped, sequences of substantially equal length and having predetermined autocorrelation values. The sequences may resemble noise in at least some of their characteristics. The orthogonality or cross-correlation characteristics, the autocorrelation characteristics and the resemblance to noise are due to features derived from sequences of pseudo-random numbers which themselves resemble noise in at least some of their characteristics. The waveform set based on the sequences is modulated digitally. The modulated set may be summed together along with a wideband reference signal of reduced amplitude and optionally an FM analog signal to form a composite signal which is broadcast typically through free space to at least one receiver. The receiver separates the analog FM signal from the digital signal and thereafter demodulates the digital data-carrying waveforms and outputs a stream of digital data. It has been determined to be resistant to multipath degradation.

This is a Division of application Ser. No. 08/485,599 filed Jun. 7,1995, now U.S. Pat. No. 5,745,525, which is a continuation of Ser. No.08/274,140, filed Jul. 12, 1994, now U.S. Pat. No. 5,956,624, thedisclosure of which is incorporated by reference.

BACKGROUND OF THE INVENTION

This invention relates to a method and system for simultaneouslybroadcasting and receiving digital and analog (e.g., FM) signals in amultipath environment.

In recent years, the quality of commercial audio broadcast signals asdelivered by radio transmitters through atmospheric free-space has beeneclipsed by the quality of stored program material, such as digitalcompact disc and audio tape technology. The quality differential of suchstored digital program material over conventional analog frequencymodulated (FM) broadcasting is so significant that there has been amarket shift in listener preference to the stored digital programmaterial. Further adding to this market shift is the increaseddegradation of FM signal quality, particularly in highly urban areas,due to multipath and noise.

Signals with line-of-sight propagation are subject to interference andfading from reflected copies of the signal, both narrowband andwideband. Such interference resulting from the simultaneous receptionvia multiple propagation paths between the transmitter and receiver iscommonly referred to as multipath (MP), the different propagation pathshaving varying times-of-arrival, amplitude, and phase.

One of the most difficult environments in which to achieve high qualitydigital radio communication is the mobile reception of atmospheric freespace signals in urban areas. The principal impairment in suchenvironments arises from multipath. Tall buildings and the like act asstrong MP reflectors, particularly in the very high frequency (VHF)region of from about 30-300 MHz.

The adverse effects of multipath (MP) on an isolated signal waveform maybe grouped at least into the following three categories: fading,dispersion, and intersymbol interference. Fading involves rapidamplitude variation as propagation paths constructively anddestructively interfere, but may be controlled under certaincircumstances with automatic gain control (AGC) circuits.

Dispersion is caused by time-varying phase disruption within and betweenbauds (symbols), and may be controlled under certain circumstances withan automatic equalizer.

Intersymbol interference (ISI) is caused by the interaction of onesymbol (or waveform) with other symbols in time. An automatic equalizer,which may be used to correct phase dispersion, may compensate foradverse effects of intersymbol interference when the symbol shape of theinterfering waveform is approximately identical to that of the desiredchannel. However, it is difficult and expensive to correct intersymbolinterference from undesired propagation paths which represent differentsymbols of bit patterns that substantially precede or succeed thedesired symbol in time.

Therefore, a conventional correction technique is to increase the timeduration of the symbol interval (or baud interval) to be much longerthan the expected multipath delay. Typical expected multipath delaysgenerally range from a maximum of about 5 to about 30 microseconds (μs)for a VHF channel. However, the increase in time of the baud or symbolinterval leads to a decreased data rate.

In multichannel systems with increased baud intervals, the shapes andcharacteristics of the basis waveforms have significant influences onthe BER. (Basis waveforms are the unmodulated sequences representing thedata carrier of each channel.) Accordingly, there has been research intodesirable characteristics of signal waveforms usable in suchmultichannel environments for producing superior performance. Thisresearch has often been conducted in combination with the use ofconventional correlation receivers. In a conventional correlationreceiver, a satisfactory received signal is one which satisfies thespectral confinement requirements of the particular application, and ischaracterized by predetermined crosscorrelation and autocorrelationproperties.

The cross-correlation property (or orthogonality) is measured between asingle signal waveform in a set and all other members of the waveformset. Low cross-correlation is important in multichannel carrier systemsin order to ensure that the individual carriers may be recovered andrecognized independent of one another. The cross-correlation representsthe degree to which a particular waveform is mathematically correlatedwith one or more other waveform in the set. The smaller the absolutevalue of the cross-correlation between any two waveforms, the moreunique are the waveforms in the correlation sense. Therefore, an idealsignal set for a correlation receiver has a cross-correlation of closeto about zero at the sampling point among all pairs of the set. (Inother words, it is a set where the waveforms are mutually orthogonal.)Good cross-correlation properties are also required for satisfactorychannel performance absent multipath because channels act as sources ofinterference to each other.

Good autocorrelation is of primary importance in multipath environmentsbecause reception requires distinguishing among similar signals withvarying times of arrival. (Autocorrelation is a measure of how unique asignal is when compared to itself in a correlation receiver when shiftedin time by a positive or negative amount of time shift. An ideal signalset with respect to autocorrelation is one where the autocorrelation foreach signal is at a minimum (or has a low value) for substantially allpositive and negative time shifts and is at a maximum for about zerooffset or, in other words, for relatively no time shift at all.)

Signal waveforms-constructed from amplitude samples of unconstrained (orunshaped) and non-orthogonal noise sequences have been proposed andutilized in prior art communication systems (e.g., spread spectrumapplications). In a similar manner, prior art systems have utilizedprime polynomials to generate pseudo-random binary sequences (also knownas PN or direct sequence) which are limited to the values +1 and -1.Such bi-valued systems possess noise-like properties to a limitedextent.

The prior art method of Code Division Multiple Access (CDMA) utilizeslong baud intervals in a plurality of digital data channels, eachcarrier being a binary sequence obtained from, for example, Gold codesor Rademacher-Walsh codes. CDMA systems are spread spectrum systems thatuse multiple binary-valued codes to achieve a higher throughput orincreased capacity than a single spread spectrum code. CDMA codesgenerally must make a tradeoff between cross-correlation andautocorrelation, but typically cannot satisfy acceptable characteristicswith respect to both.

A primary disadvantage of CDMA is that it does not permit spectralshaping of the carrier(s) without significant destruction of thesequence properties. Additionally, the number of different acceptablesignals which may be generated by CDMA codes is limited by the bi-valuednature of such signals.

In applications where spectrum compliance is not an issue,direct-sequence spread spectrum techniques which utilize noise-likewaveforms are effective in combating multipath. However, existingtechniques for constructing noise (or the more restrictive example ofpseudo-noise) waveforms do not permit arbitrary constraints in the shapeof their spectral response without significantly disrupting theresulting waveform properties. This is important because practicalsystems require band limiting filters or similar processing in order tostay within a fixed frequency allocation and/or reject particularnarrowband interference. Furthermore, although the cross-correlation issmall in spread spectrum systems, it is generally non-zero and hence thesignal waveforms act as interferers to one another even in the absenceof multipath.

U.S. Pat. No. 5,278,826 discloses a method and apparatus for digitalaudio broadcasting and reception wherein a system is provided fortransmitting and receiving through free space a composite signalconsisting of a frequency modulated (FM) analog signal and amulticarrier modulated digital signal which is especially adapted to beresistant to multipath degradation. The FM signal and digitalmulticarrier modulated signal are fully coherent. The digital signalcomprises a plurality of carriers having a maximum amplitude at least 20dB below the unmodulated FM signal, preferably 30 dB below the FMsignal. Unfortunately, the multicarriers making up the digital signal inthis patent are narrowband in nature, each carrier or channel being asingle tone which is phase modulated. A problem with such carriers isthat multipath (MP) is a frequency selective phenomenon which alters ordestroys some frequencies while letting others alone. Thus, narrowbandcarriers are extremely vulnerable to the adverse effects of multipath.Furthermore, the digital frequency spectrum in this patent is extremelyclose to the FM center frequency, thus resulting in interference betweenthe FM and digital signals.

U.S. Pat. No. 4,403,331 discloses a method and apparatus fortransmitting digital data over limited bandwidth channels, with a set ofwaveforms being mutually orthogonal to one another and bi-phase datamodulation in order to use a correlation-type multiple channel ormulticarrier receiver. This patent discloses a technique for determiningeigenvectors for the basis functions which maximize the spectraloccupancy of the carrier waveforms primarily by utilizing a longer baudinterval. The basis functions are based on a fixed number of sinusoids(which are not noise-like), and the system utilizes optimization in thefrequency domain. Unfortunately, this does not translate into goodautocorrelation properties or result in waveforms which may be madephase-continuous at the baud boundaries. The lack of phase continuity atbaud boundaries increases intersymbol interference, thereby limiting theability to properly receive signals with good BER. Optimization in thefrequency domain does not translate necessarily into optimization in thetime domain.

SUMMARY OF THE INVENTION

According to the invention, a system and method for transmitting digitalinformation through a medium such as atmospheric free-space includes atransmitter which generates a signal from a wideband spectrally-shapedset of mutually orthogonal basis signal waveforms of substantially equalduration and bandwidth and which have desired autocorrelationcharacteristics. The sequences may resemble noise in at least some oftheir characteristics. The orthogonality or crosscorrelationcharacteristics, the autocorrelation characteristics and the resemblanceto noise are due to features derived from sequences of pseudo-randomnumbers which resemble noise in at least some of their characteristics.The waveform set or corresponding digital sequence set is modulateddigitally and is resistant to multipath degradation. Digitally modulatedorthogonal waveforms making up the modulated set may be summed togetheralong with a wideband reference signal of reduced amplitude andoptionally with an FM analog signal to form a composite signal which isbroadcast typically through free space to at least one receiver. Thereceiver separates the analog FM signal from the digital signal andthereafter demodulates the digital data-carrying waveforms and outputs astream of digital data. An FM radio station may simultaneously broadcastthe same program both digitally and via analog FM using the samecomposite signal and transmitting antenna.

A basis signal set is provided for use in developing the broadcastsignal. The basis signal set comprises a plurality of different widebandwaveforms which have at least some characteristics resembling noise, theset being constructed of numerical sequences having characteristicsresembling noise but which are of a predetermined length, havepredetermined autocorrelation characteristics and wherein the pluralityof wideband sequences making up the signal set are substantiallyorthogonal in pairs as referenced to a sampling point in time.

A line-of-sight (FM broadcast band) wideband signal is robust tomultipath degradation and intersymbol interference. The inventionprovides a high spectral efficiency digital communication link inchannels with known colored interference and multipath.

This invention will now be described with respect to certain embodimentsthereof, accompanied by certain illustrations.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a broadcast system including a transmitterand receiver according to a first embodiment of this invention.

FIG. 2 is a block diagram of the transmitter portion of the firstembodiment of this invention depicted in FIG. 1.

FIGS. 3A and 3B are a schematic illustration of certain advantagesassociated with long baud intervals in multipath environments.

FIGS. 4A and 4B are graphs illustrating narrowband and wideband(broadband) signal sets, respectively, with respect to frequencyallocation.

FIGS. 5A, 5B, 5C, and 5D are a plurality of graphs illustrating theimpact of multipath nulls or interference upon both narrowband andwideband signal sets.

FIGS. 6A and 6B are graphs illustrating the adverse effects of coloredinterference upon narrowband and wideband carriers or signal sets,respectively.

FIG. 7 is a flow chart illustrating steps utilized in generating awideband orthogonal signal set having good autocorrelationcharacteristics according to the first embodiment of this invention.

FIG. 8 is a graph illustrating five exemplary random number sequencesresembling noise as generated in the first embodiment of this invention.

FIG. 9 is a graph illustrating the shaping function unit impulseresponse used to shape the random number sequences of FIG. 8 accordingto the first embodiment.

FIG. 10 is a graph illustrating the frequency response of the shapingfunction of FIG. 9 according to the first embodiment.

FIG. 11 is a graph illustrating the random number sequences of FIG. 8after spectral shaping by the impulse response of FIG. 9 according tothe first embodiment.

FIG. 12 is a graph illustrating the frequency response of the spectrallyshaped sequences of FIG. 11 according to the first embodiment.

FIG. 13 is a graph illustrating the crosscorrelation or orthogonalitycharacteristics of the spectrally shaped sequences or waveforms of FIGS.11-12 according to the first embodiment.

FIG. 14 is a graph illustrating the autocorrelation characteristics ofthe spectrally shaped sequences of FIGS. 11-12 according to the firstembodiment.

FIG. 15 is a graph illustrating the determined spectrally shaped,orthogonal sequences or waveforms after SVD matrix decompositionaccording to the first embodiment.

FIG. 16 is a graph illustrating the frequency response of the sequencesor waveforms of FIG. 15 after SVD decomposition according to the firstembodiment.

FIG. 17 is a graph illustrating the cross correlation characteristics ofthe sequences or waveforms of FIGS. 15-16 according to the firstembodiment.

FIG. 18 is a graph illustrating the autocorrelation characteristics ofthe sequences or waveforms of FIGS. 15-17 according to the firstembodiment.

FIG. 19A is a spectral diagram illustrating the spectral allocation ofthe digital and analog FM signals as linearly summed and transmittedover atmospheric free space according to the first embodiment.

FIG. 19B is a block diagram of a portion of the transmitter according toa second embodiment.

FIG. 19C is a block diagram of another portion of the FIG. 19Btransmitter.

FIG. 20 is a block diagram illustrating the receiver of the firstembodiment of this invention.

FIG. 21A is a block diagram/flowchart of the baud clock controller ofthe receiver for the first embodiment illustrated in FIG. 20B.

FIG. 21B is a flowchart for the baud clock controller of FIG. 21A.

FIG. 21C is a flowchart illustrating the steps taken for carrier clockrecovery in the receiver illustrated in FIGS. 20A and 20B.

FIG. 21D is a block diagram of certain signal processing units of thereceiver depicted in FIGS. 20A and 20B according to a particularhardware embodiment of this invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now more particularly to the accompanying drawings in whichlike reference numerals indicate like parts.

In FIG. 1, a system for simultaneously broadcasting and receivingdigital and analog signals invention includes transmitter 1 and receiver3. Transmitter 1 simultaneously transmits or broadcasts both an analogfrequency modulated (FM) signal and a digital signal, the two signalsbeing summed together in summer 9 to form a composite FM/digital signal.Preferably, when used for broadcast to automobiles and the like, thedigital signal and the analog FM signal correspond in their content whendemodulated by receiver for listening by a user, although this need notbe the case. Transmitter 1 includes digital signal generator 5 andanalog FM signal generator 7.

The composite FM/digital signal broadcast by transmitter 1 may bereceived by either a conventional analog FM receiver, a digitalreceiver, or a combination FM/digital receiver 3 as illustrated in FIG.1.

Upon receiving the composite FM/digital signal at the radio frequency(RF) carrier frequency via an omnidirectional antenna (not shown), thecomposite FM/digital signal is divided into two paths by way of splitter11.

A conventional analog FM demodulator 13 interprets the analog FM signaland outputs it in a conventional manner. The digital signal fromsplitter 11 is directed toward digital signal processor and demodulator15. Digital processor and demodulator 15 processes the digital portionso as to output a digital audio data stream for reproduction.

The composite signal may also be received by a conventional FM receiverhaving a conventional FM demodulator 13. Alternatively, the compositesignal may be received by a purely digital receiver according to anotherembodiment of this invention including only a digital signal processorand demodulator 15.

FIG. 2 is a more detailed block diagram of transmitter 1 shown inFIG. 1. Analog FM signal generator 7 generates an analog frequencymodulated signal in a conventional manner utilizing analog source 17,signal compositor (or audio stereo generator) 19, FM modulator (orexciter) 21, upconverter 23, local oscillator (LO) 25, and clippingamplifier 27.

Analog source 17 may be any conventional program source but is typicallyan audio program source with the possible addition of a data or audioprogram source on a so-called SCA side carrier. Signal compositor 19 isrepresentative of the stage of analog FM signal generation whichproduces the composite baseband signal, sometimes including a pilottone. From source 17 and compositor 19, the signal is directed tofrequency modulator or exciter 21 which is a subsystem for modulatingthe composite baseband signal of compositor 19 in a frequency typicallyacross about a 200 kHz bandwidth or less in proportion to instantaneousamplitude of the composite signal to produce an output signal centeredabout a center frequency.

After signal processing via frequency modulator 21, the modulated signalis at an intermediate frequency (IF) and is then upconverted tothe-desired output frequency by upconverter 23 and local oscillator (LO)25. The FM signal is thus centered about the center frequency of aparticular broadcast channel. The upconverted signal may then be applieddirectly to summer 9 and transmitting antenna 29 for radiation, oralternatively may be processed through clipping amplifier 27.

Alternatively, upconverter 23 and LO 25 may be eliminated and in such acase exciter 21 would directly generate the final FM signal at the RFfrequency.

FM modulator or exciter 21 may be a voltage-controlled oscillator (VCO)whose modulation is controlled by compositor 19. Furthermore, an SCAgenerator (not shown) may also feed exciter 21 in certain embodiments.

Digital signal generator 5 of transmitter 1 outputs a multicarrier ormultichannel digital data signal to signal summer 9 using a digitalprogram source 31, a digital compression subsystem 33, an errorcorrection encoder 35, a basis sequence or waveform generator 37, adigital data modulator/multiplier 39, a reference signal generator 41, amodulated set and reference signal summer 43, a digital-to-analog (D/A)converter 45, an upconverter 47, a local oscillator 49, and a linearamplifier 51. Separate amplifiers are utilized for the digital andanalog FM signals because the digital signal power level is smaller thanthe FM power level in certain embodiments, so a clipping amplifier canbe used for the FM signal.

Alternatively, use of an IF may be avoided in the transmitter as shownin the transmitter embodiment of FIGS. 19B-19C hereinafter described. Inthis embodiment, the signal is directly modulated at the final RFcarrier frequency by sending the samples from the MAC to an In-phasechannel (I) D/A converter and a Quadrature-phase channel (Q) D/Aconverter in alternating fashion and using a hybrid I/Q modulator.

With reference again to FIG. 2, digital program source 31 may produceany conventional digitized program material, such as digitized audiodata. Coupled to digital source 31 is digital compression subsystem 33which reduces the bandwidth requirements of the audio signal. A suitablecompression subsystem may, for example, operate in accordance with theMUSICAM standard described in U.S. Pat. No. 4,972,484. However, othercompression is envisioned

The compressed digital audio data making up a digital data stream ofabout 256 kbit/sec for example is then sent to error correction encoder35 which adds redundancy of, for example, up to 128 kbit of errorcorrection code in order to assist in signal recovery. Thus, the digitaldata stream emitted from error correction encoder 35 is the combinationof the 256 kbit/sec data stream input into encoder 35 and the 128 kbitof error correction code to make up about a 384 kbit/sec data stream tobe forwarded to modulator/multiplier 39. Encoder 35 may include a datascrambler which ensures that the resulting binary sequence issubstantially random in pattern.

In certain embodiments of this invention, the audio signal is protectedat error correction encoder 35 by a combination of different errorcorrection codes. For example, the central audio portion or main portion(or other critical synchronization data) of the signal may be protectedby a Viterbi code which works at high levels of interference and isfairly aggressive in nature, while the higher and lower portions of thesignal are encoded with Reed-Solomon block coding. Thus, a combinationof Viterbi and Reed-Solomon error correction codes may be used toprotect the audio signal, with each code protecting a different portionof the signal. Apparatus for such a combination of Viterbi andReed/Solomon encoding may be obtained from Corporate Computer Systems(CCS) based in Holmdel, N.J. Alternatively, conventional Reed-Solomonerror correction code may be utilized to protect the entire audiosignal.

When the combined audio and error correction code (ECC) data proceedingat a rate of 384 kbit/sec reaches modulator/multiplier 39, it is appliedto a set of orthogonal basis sequences or waveforms from generator 37.

FIGS. 3-18 herein are used to describe the functionality and output ofbasis sequence generator 37. FIGS. 3-6 illustrate certain advantages ofwideband signals, and FIGS. 7-18 illustrate the characteristics andprocesses utilized for generation of the set of basis sequences orwaveforms. The generator 37 in this example produces a set of 48 uniquesequences or "waveforms" for each baud, each corresponding to adifferent digital carrier or channel.

FIGS. 3A and 3B illustrate two approaches for digital data transmission,FIG. 3A illustrating a multipath delay greater than the baud interval orlength being utilized, and FIG. 3B illustrating a lengthened baudinterval resulting in the multipath delay being shorter than the baudinterval. In FIG. 3A, the relatively short baud interval causes baud #1as delayed to overlap baud #2 as directly transmitted, the delayed baudpattern being illustrated below the directly transmitted baud patterunaffected by multipath. Because the baud interval in FIG. 3A is shorterthan the multipath delay, substantial intersymbol interference results.

However, as illustrated in FIG. 3B, because the baud interval is longerthan experienced multipath delay, intersymbol interference is reducedbecause delayed baud #1 only overlaps directly transmitted baud #1 (notbaud #2). Thus, low baud rates (equivalent to longer baud intervals) aremore robust or resistant to multipath problems, the baud rate beingdefined as the reciprocal of the baud period or interval. There is alimit to the length of a baud--if the baud is too long, then the carrierfrequency may not be sufficiently stable during the entire baudinterval, particularly in mobile situations where the Doppler frequencyvaries as a vehicle containing the receiver changes velocity. Thetypical practical limit is less than about a millisecond.

Unfortunately, long baud intervals reduce intersymbol interference atthe expense of data rate. The use of baud intervals longer than theexpected multipath delay keeps the transmitted bauds synchronized andreduces error due to intersymbol interference but results in lower datarates than do the shorter baud times shown in FIG. 3A. Therefore, toachieve the same data rates in the systems of FIGS. 3A and 3B, thesystem utilizing longer baud times (e.g., FIG. 3B) must increase thenumber of transmission channels or carriers within a particularbandwidth (i.e., more channels or carrier per baud are required).

Wideband signals reduce the impact of multipath (MP) induced fading.FIGS. 4A and 4B illustrate and compare a narrowband (e.g., COFDM) set ofN channels and a wideband set of N channels respectively.

In the narrowband case of FIG. 4A, each channel or carrier is assigned aparticular and substantially different frequency, and each channel orcarrier is functionally separated from the other channels at thereceiver by a frequency filter. Thus, in the narrowband case illustratedin FIG. 4A, channel 1 is at the lowest frequency while channel N is atthe highest frequency. In contrast to the narrowband approach, signalswithin a wideband set have the same general frequency characteristicsand cover substantially the entire available frequency band. Thus, inthe wideband case shown (in three dimensions) in FIG. 4B, channel 1 andchannel N both cover substantially the entire band.

Wideband waveforms which define these signals have two significantrequirements. The signals must be designed so that they are orthogonalin nature at the sample point, thus rendering them separable by amatched filter or MAC within the receiver, and the spectral shape of thetransmitted signals must be such as to avoid conflict with coloredinterference. The advantage of wideband signals in combatting multipathnulls is illustrated in FIGS. 5A and 5B, which are graphs illustratingthe response of channels or carriers without multipath nulls, while thegraphs of FIGS. 5C and SD illustrate channels experiencing multipath(MP) nulls.

Multipath is considered a frequency specific interference, so themultipath null shown in the graphs of FIGS. 5C and 5D completelydestroys one or more narrowband carriers or channels, and severelyattenuates several others, but it does not completely destroy anywideband carriers. Therefore, in the face of a MP null, the error ratefor a narrowband system cannot be better than 1/N, where N is the numberof channels.

The multipath null in the wideband channels of FIG. 5D impacts allchannels to a certain extent, but it leaves them all substantiallyintact. Each wideband channel thus has an opportunity to survive due toits remaining power.

FIGS. 6A and 6B illustrate the impact of colored interference CI onmultichannel or multicarrier receivers for receiving both the narrowbandand wideband type signals. FIG. 6A illustrates narrowband signal impact,and FIG. 6B depicts wideband signal impact. Colored interference isinterference with a strong power at one frequency region (covering oneor more frequencies), and weak power at other frequency regions, that isinterference having an unequal distribution of interfering energy as afunction of frequency.

Because the wideband signals S1-SN illustrated in FIG. 6B all havesubstantially the same spectral shape and cover the entire frequencybandwidth, the impact of colored interference is minimized because eachsignal is left with an ample amount of power, even though all widebandsignals suffer at least some impact due to the colored interference.

However, in the case of the frequency specific narrowband channels, thecolored interference illustrated in FIG. 6A completely overwhelmsnarrowband channels 1-3, but it has no impact on channel N.

Baud lengths or intervals of about four times the longest expectedmultipath delay may be used in certain embodiments of this invention.Due to the numerous advantages discussed above with respect to widebandsignal sets, basis sequence/waveform generator 37 generates a basissignal set made up of a plurality of wideband signals, each signal ofthe set having the same general frequency characteristics and coveringsubstantially the entire available frequency band.

The basis set generated by generator 37 should be substantiallyorthogonal (or have a crosscorrelation of close to about zero at thesampling point of a waveform), so that each of the wideband waveformscan simultaneously coexist without interfering with one another.

Additionally, the basis set should have good autocorrelation propertiesfor satisfactory performance in multipath environments, becausereception in the face of multipath requires distinguishing among similarsignals with varying times of arrival. An ideal signal set with respectto autocorrelation is one where each waveform or carrier within the sethas an autocorrelation value of close to about zero for substantiallyall positive and negative time shifts, and an autocorrelation valuewhich is at a maximum for no shift at all (or at zero time shift).

Such characteristics must be maintained in situations where thewaveforms derived from the basis set are spectrally shaped with respectto frequency. Thus, the basis signal set output by generator 37 includesa plurality (e.g., 48 for each baud) of sequences which are mutuallyorthogonal, have good autocorrelation, and occupy a finite signalbandwidth (i.e., are spectrally shaped).

FIG. 7 is a flow chart illustrating the steps taken by generator 37 ingenerating the basis sets.

Generation begins with the numerical generation of a set of randomnumber sequences with zero mean and uniform variance (Step 53). Thechoice of probability density function (PDF) for the random numbers isnot critical. Uniform and normal PDFs have been found to havesatisfactory performance for composite analog FM and digital audio data.The random numbers may be generated by a variety of conventionalalgorithms, e.g., shuffled linear congruential or subtractive algorithmsas for example set forth in Numerical Recipes: The Art of ScientificComputing., pp. 192-199. Step 53 produces uncorrelated numbers which arein nature across the entire set of sequences generated. Graphs of fiveexemplary such sample sequences depicted as waveforms are shown in FIG.8. The maximum length of each sequence of random numbers is determinedby the ratio of the baud interval to the sample interval. For example,the maximum length may be equivalent to 192 samples. However, as aresult of further processing, it is often desirable that the sequencelength of each waveform be further reduced by approximately the lengthof shaping functions hereinafter explained. In such a case, the randomnumber sequences may have about 144 samples each. The number ofsequences all of which resemble noise generated in step 53 must be atleast as great as the number of data carrying elements required per baudin the overall transmission/reception system. In certain embodiments,twice as many sequences as the desired number of data carrying elementsmay be created. For example, where there is a data rate requirement ofabout 384 kbit/sec at a baud rate of about 8 kbit/sec, at least about 48data carrying elements are required. Therefore, from about 48-100 randomnumber sequences are generated in certain embodiments.

Unlike CDMA, with its two-valued coding scheme, according to theinvention the sequences may take on significantly more distinguishablecode values because multivalued random number generation with zero meanand uniform variance results in waveforms, which according to one aspectof the invention resemble noise and are therefore referred to as beingnoise-like, allowing greater degrees of freedom than do two-valuedcodes, and thus results in waveforms having good autocorrelationcharacteristics. The parameters "baud interval" and "sample interval"are parameters which are taken into consideration in designing thesystem according to the first embodiment of this invention. The sampleinterval (or equivalently the reciprocal of the sample rate) is theincrement of the underlying grid of time intervals on which the digitalamplitude samples are constructed and used in making up the basiswaveforms. The maximum sample interval is set by the required Nyquistbandwidth of the shaping function. Thus, the maximum sample interval isabout twice the shaping function bandwidth.

While there is no minimum sample interval, it is advantageous in certainenvironments to use as large an interval as possible, due tocomputational complexity increases associated with smaller intervals.For example, the sample rate for generating the basis waveforms may beabout 1.536 million samples per second, (MSPS) with the sample intervalbeing the reciprocal of this value (651 ns).

The "baud interval" is the interval of time which corresponds to theduration of a waveform or a symbol. Basis generator 37 outputs-the sameset of 48 carriers every increment of time defined by the baud interval.The product of the reciprocal of the baud interval and the number ofdata carrying elements herein loosely referred to as channels orcarriers is the overall throughput (e.g., 384 kbit/sec) of thecommunication system. Increasing the baud interval improves theperformance of the system by reducing the effect of intersymbolinterference. Furthermore, the longer the baud interval, the moreeffective the method in determining sets of waveforms with the desiredautocorrelation and crosscorrelation properties. The maximum baudinterval is limited only by the receiver complexity, and in certainembodiments of this invention the baud interval may be about 125 μs.Thus, generator 37 in the first embodiment outputs the same basis set of48 sample-space waveforms every 125 μs when such is the baud interval.

The sequences of FIG. 8 (representing five of at least 48) are generallyunsuitable for direct use because they have poor crosscorrelationproperties and are not subject to any spectral confinement. In fact,these random number waveforms generated in Step 53 are wideband all theway to the Nyquist frequency. Therefore, each of the (48) sequences isspectrally shaped by a shaping function (Step 55). An exemplary shapingfunction is illustrated in FIGS. 9-10, with FIG. 9 showing the shapingfunction unit impulse response in the time (sample space) domain andFIG. 10 illustrating the frequency response (in the frequency domain) ofthe same shaping function. As shown, the shaping function is made up ofabout 48 samples. The shaping function is a sequence which describes theapproximate spectral confinement to which the 48 waveforms or sequencesof the basis set must adhere. This function is represented as a unitimpulse response as shown in FIG. 9, with a corresponding sampleinterval as shown (each interval of time being about 1/1.536 MHz or lessthan 800 nanoseconds between samples). The sample interval of theshaping function is the same as the sample interval for the waveforms incertain embodiments. The length of the shaping function varies accordingto the complexity of the desired bandwidth shaping requirements. Theshaping function may be determined by way of a variety of numericaltechniques such as, e.g., the algorithms of frequency sampling, bilineartransformation, and equiripple approximation. The shaping function shownin FIGS. 9-10 was determined by the method of frequency sampling.

A significant characteristic of the frequency response of the shapingfunction as shown in FIG. 10 is a region of attenuation or null betweentwo separate and spaced-apart passbands 57 and 59. Passbands 57 and 59are centered approximately 150 kHz away from a spectral center of 384kHz, each passband being from about 75-150 kHz wide, most preferablyabout 100 kHz wide with tails out to about 150 kHz. Thus, each passbandmay extend from about 100 kHz away from the center frequency to about250 kHz away from the center frequency (See FIG. 19A). The null betweenpassbands 57 and 59 of the shaping function provides a location for theanalog FM signal carrier. Alternatively, any type of known coloredinterference may be disposed between lobes 57 and 59. This void regionbetween passbands 57 and 59 permits the rejection or extraction of theanalog FM interference under modulation centered about the centerfrequency by the digital processing portion of the receiver. In analogFM applications, substantially the entire analog signal is disposedbetween and spaced from passbands 57 and 59.

According to the invention, each sequence of the set of 48 (FIG. 8) ismathematically convolved with the shaping function unit impulse sequence(FIG. 9) in order to generate the spectrally shaped set of 48 sequences,five of which are represented as waveforms in sample space in FIG. 11.This convolving function is a point-by-point multiply and add typeprocess. The spectra with respect to frequency of five of the now shapedrandom number sequences is roughly depicted in FIG. 12 (a computer plotbeing necessary to accurately reproduce the actual shape). While the 48(five of which are shown) sequences or waveforms now satisfy desiredspectral compliance, they still have poor crosscorrelation properties.In other words, after spectral shaping of the FIG. 8 sequences, thesequences ate not mutually orthogonal, although they do have goodautocorrelation values.

FIG. 13 depicts in sample space a series of waveforms showing less thandesirable crosscorrelation characteristics of the spectrally shapedsequences of FIGS. 11-12. If these signals were orthogonal (which theyare not), all crosscorrelation terms arising from waveforms other than asingle one of interest would be about zero at a specific sampling pointThis clearly is not the case in FIG. 13.

FIG. 14 illustrates the satisfactory autocorrelation characteristics ofthe five of the 48 individual sequences shown in FIGS. 11-12. Asillustrated, the shaped sequences have autocorrelation values which areat a maximum near about zero offset, and at a minimum at substantiallyall positive and negative time shifts. Thus, the now spectrally shapedsequences must be improved with respect to their crosscorrelationvalues.

In order to improve the crosscorrelation characteristics of thespectrally shaped waveforms, five of which are shown in FIGS. 11-12, thesequences or waveforms are decomposed by way of a process calledsingular value decomposition (SVD). This process, also known asPrincipal Components Analysis, is described in Numerical Recipes: TheArt of Scientific Computing by William H. Press, copyright 1986, pp.52-64, the disclosure of which is hereby incorporated herein byreference, but which is explained herein. SVD is based upon the theoremthat any M x N matrix A, whose number of rows M is greater than or equalto its number of columns N, can be written as the product of:

an M×N matrix U,

an N×N diagonal matrix W with positive or zero elements, and

the transpose of an N×N orthogonal matrix V,

with matrices U and V being column-orthonormal. The shapes of thesematrices are illustrated as follows: ##EQU1## The SVD process isperformed on an arbitrary matrix A, replacing it by U and returningmatrices W and V separately. A preferred computer program to performthis transformation, known as SVDCMP, is based on a program by Forsytheet al. which is in turn based on the original routine of Golub andReinsch found in various forms in Wilkinson and Reinsch, in UNPACK, andelsewhere. These references, which are readily available to those ofskill in the art, include extensive discussions of the algorithm. Inapplying the SVD process to the spectrally shaped waveforms of FIG. 11(and the other 43 waveforms), an oversampled or overdetermined matrix Ais constructed by arranging the spectrally shaped waveforms generated bygenerator 37 as columns of the matrix A, matrix A having M rows and Ncolumns where the number of rows M generally being greater than thenumber of columns N (Step 57, FIG. 7). Accordingly, the number of rows Mrepresents the number of samples in a baud (or sequence), and the numberof columns N represents the number of sequences or waveforms. Afterconstructing matrix A in such a manner, the matrix is decomposed via SVDinto the product of three additional matrices: U, W, and V (Step 59).

    A.sub.M,N =U.sub.M,N W.sub.N,N V.sub.N,N.sup.T

where ##EQU2##

The decomposition of matrix A by using SVD creates matrix U whichpossesses M rows and N columns (matrix U is the same size as matrix A),wherein the columns of matrix U are orthogonal in pairs. In other words,the left-most 48 columns of newly formed matrix U represent the basisset of sequences or waveforms in the first embodiment, these sequencesall being substantially mutually orthogonal. Furthermore, each column ofmatrix U has unit vector norm (orthonormal).

The left-most 48 columns of matrix U are sequences that represent thebasis set of desired waveforms being mutually orthogonal, spectrallyshaped, and having good autocorrelation values. Additional matrices Wand V may be disregarded. The crosscorrelation/autocorrelation ratio ofthese waveforms is less than about 0.003, and more preferably less thanabout 0.001. Because the number of columns N in matrix u is typicallygreater than the number of waveforms or channels (e.g., 48) required forthe basis set, only as many columns as needed are selected, startingwith the left-most column of matrix U and proceeding to the right. Inthe event that there are insufficient degrees of freedom due to a smalltime-bandwidth product, some waveforms, particularly those toward theright-most column of matrix U, may not satisfy the spectral confinementrequirement of the desired application. In such a case, it may benecessary to alter the shaping function or increase the baud interval.

FIG. 15 illustrates five of the spectrally shaped sequences of FIG. 11after they have been decomposed via SVD as described above. The set of48 wideband waveforms are orthogonal, spectrally shaped, and have goodautocorrelation properties. The frequency responses for the fivewaveforms of FIG. 15 are illustrated in the graph of FIG. 16. As shown,the decomposition of the waveforms via SVD, while providing goodcrosscorrelation characteristics to the basis waveform set, did notadversely affect its spectral or autocorrelation characteristics.

FIG. 17 illustrates the desired crosscorrelation (or orthogonality)characteristics according to the invention of the basis SVD decomposedsequences, illustrated as waveforms, for the sequences shown in samplespace and in the frequency domain of FIGS. 15-16. Significantly, thecrosscorrelation sequences at the sampling point 192 are all near zero,thereby illustrating the orthogonal properties of the sequences of FIGS.15-16. While the 48 waveforms in the output from generator 37 aresubstantially orthogonal at the sampling point, they often are notperfectly orthogonal due to noise and limited A/D converter resolutionin practical embodiments. As shown, for wideband signals, thecrosscorrelation can only be near zero near the sampling point and notat all points in time because of the common frequencies occupied by thewideband signals or carriers.

FIG. 18 illustrates the autocorrelation values of the five sequencesillustrated in FIG. 15. The values are excellent in that theautocorrelation is at a minimum for both positive and negative timeshifts and at a maximum for about zero offset. Thus, it has surprisinglybeen found that the application of singular value decomposition (SVD) tospectrally shaped nonorthogonal waveforms such as in FIG. 11 results ina basis set of sequences which are substantially orthogonal while thespectral shaping and autocorrelation values are maintained throughoutthe application of SVD. Although SVD has been used in the past in thesolution of least squares problems, it-has never to the best of theinstant inventors' knowledge been utilized for determining signalwaveforms in data communication applications.

The crosscorrelation/autocorrelation ratio may be used to define good orsatisfactory autocorrelation and crosscorrelation. This ratio hereinrelates to the performance of channel(s) in realistic hardwareimplementations because while the mathematical crosscorrelationcharacteristics are perfect, they are typically physically unrealizablebecause they would require infinite precision to satisfy suchperfection. In an embodiment of this invention, for BPSK datamodulation, the probability of error is (reference: Lindsey, W. C., andM. K. Simon, Telecommunication Systems Engineering, 1973): ##EQU3##where E is the bit energy and N is the noise energy. In certainembodiments herein, the measured P is less than about 1×10⁻⁸ in theabsence of noise other than the channel-to-channel interference. Throughiterative solution of the equation, E/N is thus found to be no less than16. E may be considered the peak autocorrelation sum of any basiswaveform (its energy), while N is the additive crosscorrelationcontributions of each of the remaining 47 basis sequences which areacting as sources of interference. Hence thecrosscorrelation/autocorrelation (or cross/auto) ratio is less than1/(16*47) or 0.001 for each waveform pair. In certain embodiments, a biterror rate of better than 1×10⁻⁴ is maintained, this corresponding to amaximum cross/auto ratio of about 0.003. Thus, the systems according tocertain embodiments of this invention have cross/auto ratios of lessthan about 0.003 and preferably less than about 0.001. Thecrosscorrelation characteristics typically dominate channel performancebecause the 48 channels act as source of interference with respect toeach other. In the absence of MP, autocorrelation is not as important,autocorrelation primarily reflecting the "whiteness" or broadness of thefrequency content of the basis waveforms.

FIGS. 7-18 illustrate the generation of a basis wideband waveform set bybasis sequence or waveform generator 37. Without further encoding, the48 basis waveforms of each set representing different symbols would notvary from baud to baud, hence only a constant bit sequence would becommunicated because the same set of 48 waveforms are output each baud.Thus, the basis waveform shapes making up the set must be modified ormodulated in a data dependent manner for use in the communication systemof the first embodiment.

Modulator/multiplier 39 of FIG. 2 modulates the incoming basis waveformsoutput by generator 37 in a data dependent manner in accordance with thedigital data coming in at a rate of 384 kbit/sec from error correctionencoder 35.

In order to maintain the desired properties of crosscorrelation,autocorrelation, and spectral shape, variable gain modulation, includingthe special cases of inversion (generally considered as phasemodulation) and infinite attenuation, is utilized by modulator 39. Thus,modulator 39 may apply, for example, binary phase-shift-keying (BPSK) orM-ary orthogonal modulation to the basis waveforms in certainembodiments. For the two-level case of phase-shift-keying (PSK or BPSK),one state represents the binary "1" and the other state a binary "0."For multilevel or M-ary systems, there are more than two levels orstates, usually a multiple of two, with a few exceptions such as partialresponse systems, duo-binary being an example.

Phase modulation such as PSK or BPSK uses one phase of the carrierfrequency or waveform for one binary state, and the other phase for thesecond binary state. The two phases are about 180 degrees apart incertain embodiments, and are detected at the receiver by a synchronousdetector using a reference signal at the receiver which is of knownphase with respect to the incoming signal.

Alternatively, offset quaternary phase-shift keying (OQPSK) modulationmay be performed at modulator 39. OQPSK is based upon the 90 degreecarrier phase 4 separation of two sets of sequences, so the twosequences are interleaved in time. One set of sequences is denoted asthe in-phase set, and the remaining sequences are denoted as thequadrature set. When OQPSK is used, a reduced set (e.g., 24) of basissequences is generated by generator 37 by considering only one-half thespectrum. Hence, the spectral shaping function is considered symmetricabout a midpoint, which is translated to zero frequency. For example, incertain embodiments, the shaping function is degenerated to a singlepassband 100 kHz wide and centered about 150 kHz away from zerofrequency. The in-phase waveforms are defined from the basis setwaveforms, and the quadrature waveforms are constructed bytime-reversing the in-phase waveforms. However, they may also beidentical to the in-phase waveforms.

In the first embodiment of this invention, waveform generator 37 outputsan orthogonal basis set of 48 waveforms or sequences to modulator 39 ata baud rate of about 8 kbit/sec, each sequence or waveform of the setbeing unique and representing a different channel or carrier. Atmodulator 39, the 48 waveforms are each individually multiplied byeither a positive 1 (+1) or a negative 1 (-1) in the bi-valued phasemodulation process, a sequence multiplied by +1 representing a binary 1and a sequence multiplied by -1 representing a binary 0.

After being modulated in modulator 39 so that each waveform in the setrepresents either a binary 1 or 0, all 48 waveforms making up the baudare combined at summer 43 with a wideband reference signal to bedescribed below. Thereafter, generator 37 again outputs the same 48basis sequences to modulator 39, which modulates this newly sent baud inthe same manner. Thus, a substantially continuous flow of bauds (eachbaud including 48 data channels or carriers) is output from modulator 39to summer 43, with the 48 signals of each baud along with a referencesignal from generator 41 being combined and thereafter forwarded todigital-to-analog converter 45.

When modulator 39 of the first embodiment receives a waveform fromgenerator 37 and is instructed by the incoming digital data to modulatethis waveform to represent a binary zero, modulator 39 multiplies all192 samples of this waveform by -1 and thereafter sends the modulatedwaveform to summer 43. Thus, the actual shapes of the waveforms receivedfrom generator 37 are not changed in modulator 39, only their polarityis changed.

Once a satisfactory set of 48 waveforms has been determined (by SVD),the set is stored in generator 37 and is repeatedly generated andoutput. In a similar manner, reference signal generator 41 determines asatisfactory signal and repeatedly outputs the same stored widebandreference signal to summer 43 for each baud.

The wideband reference signal output by generator 41 is unmodulated, andis not related to the data signals output from modulator 39. Thereference signal is a waveform which also possesses characteristicsresembling noise, being composed of the summation of all of the basisfunctions (e.g., 48), each basis function having either a positive ornegative polarity determined arbitrarily in certain embodiments.

The reference signal is the same for every baud, and satisfies thebandwidth occupancy determined by the system. The reference signal isscaled down in amplitude with respect to the modulated waveforms of eachbaud. To generate the reference signal, an arbitrary binary value ispicked for each of the 48 basis waveforms in the particular set outputfrom generator 37, these modulated waveforms then being summed togetherto represent a composite signal. The composite signal is then attenuatedso as to represent from about one-quarter to one-half the amplitude ofeach of the 48 signals emitted from modulator 39. The crosscorrelationof the reference signal is about 1/48 for any one waveform in the signalset.

There is a trade-off associated with reducing the amplitude of thereference signal. Increasing reference signal power reduces the numberof averages required in the receiver which thereby makes it possible touse a faster equalizer. The nominal trade-off in certain embodiments isabout 50 averages and about one-half reference signal power or amplitudewith respect to the amplitude of the modulated waveforms.

Because multipath (MP) and colored interference may impact some channelsmore than others, the reference signal is composed of all 48 channels.The reference signal is constant in every baud, so that it must besubtracted from each baud at the receiver before the demodulation stepis performed.

While the selection of polarity for each channel or carrier in making upthe reference signal is somewhat arbitrary when BPSK modulation is used,such is not the case with M-ary or on-off modulation. The compositedigital signal output from summer 43 is converted to analog by D/Aconverter 45. Alternatively, 48 separate and distinct digital-to-analogconverters may be disposed immediately after modulator 39, and summer 43would then be of the analog type.

The signal output from D/A converter 45 is at an intermediate frequency(IF) of about 384 kHz in the first embodiment. The frequency of thesignal is moved up to the desired final FM frequency by way ofupconverter 47 and local oscillator (LO) 49. This upconversion is ananalog process performed on the combined 48 data channels and referencesignal of each baud. After being upconverted, the signal is linearlyamplified via amplifier 51, and forwarded to FM and digital signalsummer 9.

Signal summer 9 linearly combines the analog FM signal output fromclipping amplifier 27 with the digital data signal output from linearamplifier 51. After being linearly combined at summer 9, the compositeFM/digital signal is sent to transmitting antenna 29 for broadcastthrough atmospheric-free space to a plurality of receivers 3.

FIG. 19A is a graph illustrating the spectral allocation of thecomposite FM/digital signal. As shown, analog FM signal 61 (or othertype of colored interference present in other embodiments) issubstantially centered about the center frequency, FM signal 61. beingsurrounded on either side by digital passband signals 63 and 65 made upof the modulated waveform set. While passbands 63 and 65 are illustratedas being substantially symmetrical, this need not be the case.

Spectral mask 67 defines the spectral limitations within which thecomposite FM/digital signal must stay, mask 67 being about 600 kHz wideand 80 dB deep in this embodiment. Mask 67 includes a central peakportion 69 and a pair of guardbands 71 immediately adjacent the centralportion. Guardbands 71 are side areas where low level conventionalanalog FM signals are typically disposed because the typical analog FMsignal cannot be attenuated in 0 bandwidth, mask 67 being a typicalanalog FM mask. As shown, guardbands 71 start out at about -25 dB frompeak power and about 100 kHz away from the center frequency.

Because typical analog FM broadcasting stations generally do not utilizeguardband portions 71 of mask 67, the digital signals of the firstembodiment are spectrally shaped so as to be disposed within theseguardband areas. Digital signals (or lobes) 63 and 65 are ofsignificantly reduced amplitude as compared to FM signal 61 due to theirpositions in the guardband areas in this embodiment, each of the 48different symbol waveforms in each baud representing portions of signals63 and 65.

While many of the functions of the aforediscussed transmitter elementsmay be carried out via the first embodiment transmitter and softwareattached hereto via microfiche, such functions may also be carried outusing hardware of a second embodiment. FIG. 19B is a hardware blockdiagram of a transmitter according to a second embodiment. The signal inthis embodiment is directly modulated at the final RF carrier frequencyby sending the samples from the MAC 191 to two D/A converters 197 inalternating succession (i.e., one for the I D/A converter, the next forthe Q D/A converter, etc.) and using a hybrid I/Q modulator.

In FIG. 19B, the input bit stream is at about 384 kHz and is acombination of MUSICAM compressed digital audio from compressor 33 andECC from encoder 35. This bit data is read into one of two 48 bit FIFO48×1 RAMs 181 and 183. The total throughput at registers 185 is about1.536 MHz which corresponds to the basis waveform sample interval. Thisthroughput is implemented with in-phase (I) and quadrature (Q) D/Aconverters (not shown) to which the samples are alternatively sent insuccession (i.e., one for the I D/A converter, one for the Q D/Aconverter, next for the I, etc.). The above-discussed basis orthogonalwaveform set of 48 different sequences is stored in ROM 187 which isindexed column by column with a modulo 9216 address sequencer 189. Thus,the first 48 matrix elements indexed correspond to the first sample ofeach basis waveform. The value of each of the 48 data bits must beaccessed 192 times (corresponding to the number of samples in a baud orwaveform) in a single baud to generate the modulated waveforms.Therefore, RAMs 181 and 183 are accessed/read in a circular fashion. Amaster crystal 6.144 MHz clock generator (not shown) is divided down togenerate the 384 kHz bit clock, the 1.536 MHz sample clock, and the 768kHz D/A converter clocks.

Modulation is done in MAC 191 which may be operated as an additive orsubtractive accumulator. MAC 191 is fast enough to complete 48 MACs in asample interval (i.e., about 1.536 MHz), otherwise multiple MACs arerequired which may be the case in certain embodiments. At the start ofeach sample interval, MAC 191 is preloaded with the value of thereferences signal for that sample, the wideband reference signaldiscussed above being stored in ROM 193 and indexed with modulo 192counter 195. The pointer to ROM 193 is reset at the start of each baudto the first sample in the ROM 193 so as to index the first value forthe reference signal.

For each of the 48 data bits in RAMs 181 and 183 corresponding to theprevious baud, if the incoming bit is a 0 the indexed value of ROM 187which represents a channel waveform sample is subtracted from thecurrent value. However, if the data bit is a 1, the indexed sample valueof ROM 187 is added to the current value. After 48 MACS, the contents ofthe accumulator represent the complete sample, i.e., the sum ofmodulated channel waveforms and reference signal. The contents are thenlatched into one of 12-bit registers 185, alternating from one sample tothe next. Registers 185 are used to drive D/A converters 197 shown inFIG. 19C.

FIG. 19C is a hardware block diagram of certain analog digital signalgenerator components which may be used according to the FIG. 19Bembodiment of the transmitter. The output of D/A converters 197 isfiltered by 400 kHz 8-pole Butterworth lowpass filters 199. D/Aconverters 197 are used to drive hybrid I/Q modulator mixer 201, whichmay be a Minicircuits MIQA100 in certain embodiments.

Local oscillator source 203 for quadrature mixer 201 is of the highstability, low phase, CW type operating at the same frequency as theanalog FM carrier. Bandpass filter 205 removes undesired sidebands andamplifier 51 is a linear power amplifier.

After being transmitted by way of antenna 29, the composite FM/digitalsignal, whose spectral allocation is shown in FIG. 19A for certainembodiments, is received by omnidirectional antenna 73 in receiver(s) 3as best shown in FIGS. 20A and 20B. The received composite FM/digitalsignal is at an RF frequency in the frequency range from about 30-300MHz and more preferably from about 88-110 MHz.

The incoming composite signal is split into two separate FM/digitalpaths, A and B, by splitter 11, the composite signal in path A beingforwarded to the analog FM signal processing section of receiver 3 andthe composite signal in path B being forwarded to the digital signalprocessing portion of the receiver.

With respect to signal path A, the analog FM signal of the composite isdownconverted to a standard intermediate frequency (IF) by downconverter75 and local oscillator 77. The output of downconverter 75 is forwardedto conventional frequency demodulator 79. The resulting signal isforwarded from demodulator 79 to conventional stereo decoder 81 foroutput via typical audio speakers 83.

Alternatively, the analog FM and digital FM portions of the receiver mayshare the same downconverter in certain embodiments, realizing, however,that the analog bandpass filter is generally more selective or narrowthan the digital bandpass filter.

Splitter 11 outputs the other FM/digital signal via signal path B towardtuned prefilter 85 which makes the first attempt at separating thedesired composite signal from interfering transmissions originating fromother stations.

The low-noise amplifier of prefilter 85 amplifies a fairly weak FMsignal level, approximately 20 dBF (dB-Femtowatts) in magnitude, to alevel of -40 dBm where it may be processed readily by further circuits.This amplifier is linear in certain embodiments so as to preserve thedigital signal.

The low-noise amplifier of prefilter 85 is followed by a high dynamicrange diode-ring mixer 87 which frequency translates the FM/digitalsignal from the RF carrier frequency to a fixed intermediate frequency(IF) of about 10.7 MHz. The choice of IF is generally dictated by theavailability of inexpensive IF bandpass filters and may, of course, varywith respect to the use of the system. Local oscillator (LO) 89 of mixer87 is sinusoidal in nature, having a frequency which is the addition ofthe IF frequency and the desired RF carrier frequency. Local oscillator89 is variable due to the requirement for tuning over the entire FMband, but once a station is selected, the frequency of LO 89 isgenerally not altered.

The signal at an IF of 10.7 MHz emitted from mixer 87 is forwarded tohigh order, 500 kHz wide bandpass filter 91 which is centered at the IFfrequency. Bandpass filter 91 separates the composite FM/digital signalfrom possible nearby adjacents so as to limit the noise bandwidth of thesystem. The bandwidth of bandpass filter 91 is significantly wider thanthat employed in conventional FM receivers because it passes the digitalsignal which surrounds the analog FM signal, the opposing lobes 63 and65 making up the digital signal in this embodiment being situated withinguardbands 71 of FM mask 67.

The bandpass filter signal is then amplified by amplifier 93 and passedthrough automatic gain control (AGC) amplifier 95 so as to present auniform signal level to the final downconversion circuits. The timeconstant of the AGC circuit, including rectifier 97 and integrator 99,must be significantly longer than the baud interval (about 125microseconds in certain embodiments), but short enough to adapt totravel transients (less than about 10 milliseconds in certainembodiments).

The composite FM/digital signal is then sent to mixer 101 which iscontrolled by frequency controlled local oscillator 103. LO 103 controlsthe carrier frequency of the receiver.

Mixer 101 frequency translates the 10.7 MHz FM/digital intermediatefrequency signal down to about a 384 kHz intermediate frequency (IF)FM/digital signal, the 384 kHz IF signal then being digitized andprocessed. The frequency of local oscillator 103 for mixer 101 is underdirect control of the receiver in order to eliminate frequency offsetsand to track Doppler frequency shifts as discussed later herein, thiscontrol being effected by focusing in on frequency shifts and not phase.

After downconversion, the signal is forwarded to triple-tuned notchfilter 105 which is centered at about the 384 kHz intermediatefrequency. Notch filter 105 substantially eliminates the analog FMcomponent of the composite FM/digital signal, the analog FM componentbeing irrelevant to reception and processing of the digital signal madeup of lobes 63 and 65. Remaining portions of the analog FM signal whichare not removed by notch filter 105 are suppressed by filtering withinlater stages of the receiver.

The resulting digital signal emitted from notch filter 105 is amplifiedby amplifier 107 to a level of about 200 mV in the absence of anyinterference, and is thereafter connected to analog-to-digital (A/D)converter 109 of receiver 3.

After the analog FM component has been removed, the signal is digitizedwith 10-bit analog-to-digital converter 109, sampling at approximately1.536 MHz. The precise frequency of A/D converter 109 is controlled bythe baseband or baud clock recovery voltage controlled oscillator withincontroller ill, the precise frequency of this oscillator/clock beingcontrolled in order to accurately track the baud frequency of thetransmitter.

The nominal amplitude presented to A/D converter 109 is chosen to allowfor a nominal dynamic range of about 20 dB with about a 15-20 dB dynamicrange overhead in order to combat adjacent channel and residual FMinterference. Over-range recovery must be less than one microsecond (As)in duration in certain embodiments so as to provide good performance inthe presence of impulse noise.

FIG. 21D is a block diagram of a hardware system which may be used toprocess and demodulate the digitized signal following digitalconversion. The digitized samples are passed from A/D converter 109through 36-tap equalizer 113 implemented as a direct-form finite impulseresponse (FIR) filter. There must be enough taps in the equalizer tospan the expected multipath delay range (5-30 microseconds for VHF).Thirty-six taps at 1.536 MHz means that the equalizer in this embodimentspans 23 As. The maximum number of taps would be the number of samplesin a baud (e.g., 192) but this usually is not practical.

The FIR filter of equalizer 113 is implemented with a fast MAC hardwareunit 114. The speed of MAC unit 114 is such that 36 MACs are performedin 1/1.536 MHz, namely the amount of time between digitized samples.Alternatively, the MAC function may be interleaved among multiple MAChardware units in order to reduce the throughput requirement for eachindividual MAC.

The delay memory for the FIR filter in equalizer 113 and the tap weightcoefficients are stored in recirculating first-in first-out RAM queues(FIFOS) 116 which have 10 bit widths. Concurrent with eachanalog-to-digital conversion at converter 109, MAC unit 114 withinequalizer 113 is cleared and preloaded, and the oldest sample from thedelay FIFO 116 is dropped. The tap weights of equalizer 113 aresequentially updated by a tap update algorithm 118 such as LMS. Othertap update algorithms such as Levinson-Durbin and Recursive LeastSquares may also be used in certain embodiments.

MAC 114 of equalizer 113 (other MAC's herein are similar) is defined asa multiply/accumulate hardware unit. It requires two input bit operands,designated X and Y, and produces a single output operand. The MAC isimplemented with parallel adders and therefore produces the outputoperand after a fixed-time interval, having been triggered by a clockpulse. The output operand is defined as the twos-complement bit-wiseproduct of the two input operands summed with the previous value of theaccumulator. The new output operand is returned to the accumulator forthe next operation.

The MAC accumulator may be cleared (reset to zero) upon the exertion ofan external clear line (not shown). Additionally, the accumulator of MAC114 may be preloaded to one of the input operands, arbitrarily chosen asthe Y operand, upon the exertion of conventional load line 120. Fordetails of a typical MAC integrated circuit, see the IDT7210L datasheets from Integrated Device Technology, Inc.,. Santa Clara, Calif.,this MAC hardware unit being used to implement filtering and correlationfunctions with the minimum width of the bit operand required varyingaccording to how the MAC is used. The operand bit widths are shown inFIG. 21D.

Tap algorithm update unit 118 is continuously updated with both theoutput of averager 117 and equalizer averager 122 in accordance with theLeast Means Square Algorithm (LMS) used by unit 118. Tap updatealgorithm 118 (FIG. 21D) and equalizer controller 118 (FIG. 20B) areequivalent.

The equalizer update algorithm makes use of the averaged referencesignal in order to determine the effect of the RF channel propagationcharacteristics. The receiver has access (i.e., stored in a ROM) to thetrue, unimpaired reference signal, and hence it may apply one of thealgorithms (e.g., LMS, Levinson-Durbin, etc.) in order to set theequalizer tap weights to linearly process the disturbed reference signalto more closely resemble the true reference signal. In the process, this"undoing" of the multipath disturbance also serves to compensate themodulated data waveforms since those signals are subject to the samepropagation disturbances as the reference signal.

In addition to being processed through equalizer 113, the digitized baudsamples are also routed from A/D converter 109 to reference signalaverager 117. Averager 117 includes a 192-element memory with an 8-bitwidth which reflects the number of digitized samples in one baud (8 kHzbaud rate at 1.536 MHz). Reference signal averager 117 takes each of the192 samples per baud and statistically averages them across multiplebauds (typically about 50 bauds) so that random variations in thesamples with respect to data waveforms are eliminated, thus leaving thebias making up the reference signal output from generator 41. Averager117 includes a single-pole infinite-impulse response (IIR) filter whichis implemented with a MAC hardware unit. The throughput of this MAC unitis modest, because it need only update once per digitized sample.

In order to facilitate operation of the equalizer tap update algorithm118 and in order to remove the bias of the reference signal from theequalized digital samples, the equalized samples are averaged with the192 element memory within averager 117. This function istime-interleaved with the same MAC unit.

The number of baud averages (e.g., about 50) is significant to properperformance, because if an insufficient number of bauds is averaged, therandom data-dependent signal will not be mitigated, thus disturbing thereference signal. In other words, if the running average of averager 117extends over too few bauds, the S/N ratio of the reference signal islow. On the other hand, if too many bauds are averaged, the S/N ratio ofthe reference signal is high and if MP changes during this averagingperiod, the averager response is greatly distorted--this compromises theperformance of the adaptive equalizer which uses the averager toestimate the RF channel's characteristics. Thus, the average is takenover from about 40-70 bauds in certain embodiments of this invention.

The contents of the averager 117 memory are directly used to determinethe tap weight updates for equalizer 113 via the EMS algorithm atsubsystem 118. The contents of averager memory 117 after being matchedfiltered at 151 are further used to determine the baud clock or basebandfrequency adjustment via controller 111, and the carrier frequencyadjustment via local oscillator 103.

The Levinson-Durbin algorithm may be used at subsystem 118 to controlequalizer 113, while the embodiment illustrated in FIG. 21D uses LMS.Alternatively, the Recursive Least Squares algorithm may be insteadused.

The baud clock frequency adjustment (also known as the baseband clockrecovery) is shown in FIGS. 20A, 20B, 21A, 21B, and 21D. The baud clockfrequency adjustment controls the 1.536 MHz A/D converter 109 samplerate via controller 111. When divided by 8, as with a 3-bit togglecounter, this frequency is also the fundamental 384 kHz bit clock.

The carrier clock frequency adjustment controls oscillator 103 viafrequency error measurer 116. Both oscillators, 103 and 111, may beimplemented with direct-digital synthesis (DDS) hardware units such asQualcomm Q2334 DDS or alternatively with phase locked loop synthesizers.At this time that there is no implied phase coherence between thecarrier clock and the baud clock--any such coherence that might exist atthe transmitter is eliminated by Doppler frequency variation and thelike.

Referring again to FIG. 21D, the equalized digital samples are loadedinto one of two 192-element RAMs 121 and 123, each RAM having a 12-bitwidth, the loading alternating from one baud to the next baud. This isrequired so that the contents of one of RAMs 121 and 123 reflecting theentire previous baud may be processed undisturbed as the equalizedsamples from the current baud are being gathered. Hence, RAMs 121 and123 operate in a "ping-pong" manner with a latency of one full baud.RAMs 121 and 123 of FIFO buffer 115 are triggered by 8-bit counters 126.

The processing required is the correlation of the equalized samples fora complete baud with each of the 48 modulated waveforms making up the 48carriers of the baud set. This processing is accomplished with MAChardware unit 127. The throughput of MAC 127 must be such that 192×48MACs are completed in the interval of one baud, this interval beingabout 125 μs in certain embodiments. RAMs 121 and 123 are implementedwithin FIFO buffer 115 shown in FIG. 20B.

Correlation with the 48 different sequences or waveforms of each baud isimplemented via MAC 127 and ROM 129. The 48 basis waveforms in each baudare stored end-to-end and organized in ROM 129 which has an 8-bit width,the address generator 131 for ROM 129 being 14-bit modulo 192×48 counter131. Thus, ROM 129 is essentially a basis sequence or waveform generatordisposed within the receiver.

MAC 127 multiplies the samples of each baud pairwise by each of the 48sequences, and then adds them up. At the start of each baud, MAC unit127 (which makes up multiplier 141 and comparator 143) is clocked 192times to accumulate the correlation of the previous baud with the firstwaveform output (e.g., "Channel 1" carrier) from ROM 129. Thereafter,MAC 127 determines the binary value of the Channel 1 carrier (i.e., 0 or1). In other words, the basis Channel 1 waveform is emitted from ROM 129and multiplied with the composite baud digital signal output from RAMs121 and/or 123 in order to obtain the modulated Channel 1 carrier.

This sample is then rounded to 3-bit precision and directed toward oneof two 48-sample RAMs 133 and 135, each with a 3-bit width. At thispoint, ROM address counter 131 will select the first sample of the nextbasis waveform from ROM 129 (i.e., Channel 2), and MAC 127 is cleared toprepare for the next correlation accumulation. This process is repeatedfor the remaining 47 sequences within the set making up one baud, eachresulting sample being sent to RAMS 133 and 135.

The 3-bit width on the digital output in RAMs 133 and 135 is only neededfor soft-decision Viterbi error decoding. If only Reed-Solomon is used,for example, the path need only be one (1) bit wide.

The 48-element RAMs 133 and 135 are organized in a ping-pong manner justas the equalized baud sample RAMS 121 and 123 are. New bit samples arewritten to one of RAMs 133 and 135 while the alternate RAM is being readby external error decoding hardware 145, such error decoding hardware145 being conventional Viterbi convolutional and/or Reed-Solomon blockdecoding hardware and algorithms. Because the binary value of eachwaveform or channel is determined by MAC 127, the digital data streamoutput of RAMs 133 and 135 is sent to error protection decoder 145 andthen to a MUSICAM decoder including a descrambler, after which it isconventional digital audio data which may be reproduced as music or thelike.

The roles of RAMs 133 and 135 alternate from baud to baud in order toallow the previous baud correlation output bit stream to be read at auniform rate while the correlation sums are produced at the MAC outputoperand. RAMs 133 and 135 as shown are indexed by address generatorsimplemented with 6-bit modulo forty-eight counters 137. Thus, multiplierand integrater unit 141 and comparator 143 made up by MAC 127, and RAMs133 and 135 simply form a correlation-type detector which reproduces bitpatterns as previously encoded at the transmitter.

FIGS. 21A and 21B are block diagram/flowcharts of the system utilizedfor controlling the baseband (or baud) of A/D converter 109via-controller 111. This system functions to establish the properfrequency of operation for the sample clock voltage controlledoscillator (VCO) in controller 111 for A/D converter 109 which must besubstantially synchronous with the baud clock in the transmitter. Thisbaud clock recovery system illustrated in FIGS. 21A and 21B alsofunctions to establish A/D sample synchronization so that the true firstsample of the baud is the first sample that is written into RAMs 121 and123 which are used to perform the matched filter correlations with MAC127.

Once the proper frequency is established, every 192 samples willcorrespond to a baud. The "baud marker" indicates which one of every 192samples corresponds to the first sample in the baud. They may be termedcontrol signals which act to clear 8 bit counters 126.

The baud clock in controller 111 is adjusted to have the same frequencyas that in the transmitter so that the data emerging from the receiveris synchronous with the transmitter data and the reference signal doesnot move relative to the baud marker. When the number of samples perbaud and the baud clock (or baseband frequency) is correct, thereference signal timing relative to the start of the baud signal willnot change. The receiver baud marker is correctly timed relative to theincoming bauds to minimize the error rate. Further, when the referencesignal does not substantially move from baud to baud, equalizer 113 cancorrectly position the reference signal to meet its timing requirementand compensate for multipath.

The digitized signal is sent from A/D converter 109 to averager 117. Inaverager 117, the data from the incoming bauds is added on a sample bysample basis with the running average of previous bauds. The dataaverages out thus leaving the reference signal. Averager 117 extractsthe reference signal from the digital signal forwarded by converter 109thus allowing averager 117 to provide a running average of the distortedreference signal. The signal-to-noise ratio of the reference signal (orchannel probe) is then maximized by matched filter 151. After beingmatch filtered, the reference signal is forwarded to both baud clockcontroller 111 and frequency controlled oscillator 103 to allow thefrequencies of these voltage controlled oscillators to be controlled.

With respect to controller 111, a moving receiver or moving multipathcauses the received FM/digital signal to be Doppler shifted, thuschanging the timing of the baud(s). However, multipath does notmaterially impact the sampling clock frequency, although it does distortthe timing. Controller 111 and the baud clock therewithin function tohold the frequency substantially constant to allow equalizer 113 toadjust the data signal (and reference signal) with respect to phase.

The reference signal should be in the same frequency band as the data tobe an effective measure of the channel. The reference signal SNR at thereceiver is increased through averager 117 because the reference signalis constant from baud to baud while noise and the data modulation israndom. Hence, the reference will coherently add while the randomperturbations will tend to cancel each other out in the averager.Therefore, it is possible to operate the reference signal at a lowertransmitter power level.

The reference signal is designed so that its position in each baud canbe measured. The processed output after averager 117 and filter 151 hasa sharp peak often in the approximate center of each baud. Becauseaverager 117 coherently adds the reference signal and incoherently addsthe data, the S/N ratio of the reference signal is increased and thefiltered version has a desired shape with a localized amplitude.

Controller 111 tracks the frequency of the filtered reference signal asopposed to its phase. It is thus blind to multipath distortion andimmune to MP-induced search caused by phase tracking. The frequencysensitive tracking of controller ill keeps the position of the referencesignal constant and allows equalizer 113 to reposition the referencesignal. MP distorts the reference signal thus causing the position ofthe largest amplitude from filter 151 to move accordingly. Controller111 measures the reference signal center of gravity (COG) as a means formeasuring and computing the precession of the reference signal relativeto the baud start sample.

The absolute value of the matched filtered reference signal is taken instep 153 (see FIG. 21B) to establish an approximate reference signalenvelope and multiplied by the time interval from the point where thebaud was assumed to begin in step 154. In other words, multiplying thesample amplitude by tie same number in step 154 relative to the firstsample in the baud produces a first moment. The products for all samplesare added together in step 156 and divided by the sum of the samples instep 158 to compute the reference signal absolute value "center ofgravity" (see step 155). Because the first moment for the matchedfiltered averager output has the same characteristics as a center ofgravity (COG) calculation, the COG technique is used for locating theeffective center of the distorted reference signal in step 155. Thus,the COG is a good way in which to compute the precession of thereference signal relative to the baud start sample.

If the baud frequency is correct, variations in distortion will causethe COG location to oscillate, but will not cause it to precess. Dopplershift of the incoming signal and baud clock frequency differencesbetween the transmitter and receiver do, however, cause precession. Theeffective error in the baud rate frequency is determined by measuringthe rate of precession of the COG location in step 157 by subtractingthe COG from the baud center. The COG for a correctly timed and MP-freereference signal is computed when the receiver is calibrated.

The rate of change of difference between the channel sampler orreference signal COG and the ideal COG is called the position error(PE). The rate of change of PE is a measure of the sample clock or baudclock frequency error. Thus, by determining the position error (PE) ofthe COG in step 157, baud clock frequency offsets and Doppler shifts aresimultaneously eliminated by adjusting the baud clock frequency in step159 via controller 111 to drive the precession rate to substantiallyzero. By determining via frequency measurement the error amount in thebaud rate in step 158, steps 170-174 adjust the voltage controlledoscillator in controller 111. If the average indicates that thefrequency is too high, a lower VCO control voltage is issued to thesample clock in step 159, and if the average indicates that thefrequency is too low, an increase in the VCO control voltage is issued.

If the PE is small, the oscillator is not adjusted (step 170) in certainembodiments. However, if there is substantial PE of the COG, step 171 iscarried out to determine the extent of the error. If it is in bounds of(or within plus or minus about 4 samples) so that equalizer 173 canhandle it, no correction signal is sent to the baseband oscillator (step172). However, if the PE is out of bounds (i.e., more than ± four (4)samples difference between the true baud center and the center as setthrough the PE algorithm), a correction signal is sent to the oscillatorin step 173, and correction is carried out in steps 174 via adjustingthe baud clock voltage controlled oscillator. It is noted that the COGis also utilized to calculate where each baud began using baud startestimator 161 thus producing the baud marker. The baud marker is a pulsewhich is output every 192 samples. A 14-bit sequencer 131 and counters126 (see FIG. 21D) both receive the baud marker pulse in order to ensuresynchronization.

The output of averager 117 and matched filter 151 is also utilized toensure correct operation of oscillator 103 thereby providing propercarrier clock frequency. FIG. 21C is a self-explanatory flowchart of thesystem for controlling oscillator 103 via the matched filter referencesignal output from filter 151. The baseband and carrier frequencies mustbe controlled by different oscillators because Doppler affects the baudclock and carrier clock to different extents. LO 103 is adjusted asfollows so the IF center frequency in the receiver is the same as in thetransmitter.

The carrier clock recovery system tracks multipath induces IF centerfrequency phase changes. The reference signal need be in the samefrequency band as the data to be an effective measure of MP distorting.

The reference signal is designed with a center frequency near (withinabout 1%) the data signal center frequency so that setting the referencesignal center frequency is substantially equivalent to setting thefrequency of the data. The IF center frequency is chosen in certainembodiments as an even harmonic of the baud rate so that the phase ofthe IF carrier with correct frequency rotates an integer×360' in onebaud period thereby allowing the receiver to interpret phase rotation ofreference signal as an IF frequency error. A frequency sensitivetracking loop keeps the phase of the reference signal constant therebyallowing equalizer 113 to reset the phase of the reference signal andalong therewith the phase of the digital data signal. As shown, timeconstant T d is defined in step 300 as about 100 times the baud periodand truncated and frequency shifted replicas F1 and F2 of the channelsampler are generated in steps 301 and 303, the channel sampler beingthe triangularly shaped weighted cosine wave output from matched filter151.

F1 is a frequency about 0.01% below the ideal IF truncated to about 10%of the baud while F2 is a frequency about 0.01% above the ideal IFtruncated to about 10% of the baud. F1 and F2 are truncated in such amanner because the reference signal is only large near the center of thebaud.

F1 is multiplied by the channel sampler on a sample by sample basis withthe products summed in step 305. The sum of products is subtracted fromthe value computed on the previous baud. The difference represents baudto baud change and frequency F1 error. This F1 error is accumulated instep 307 over several hundred bauds to get a running average F1 error.The F2 running average error is determined in steps 305 and 309 in thesame manner.

In step 311, if the F1 accumulation is equal to the F2 accumulation, LO103 is not adjusted. However, if F1 accumulation is greater than that ofF2, the oscillator frequency is increased. The opposite is true if F2 isless than F1.

The above described embodiment in which a composite signal, including ananalog FM portion and a digital portion is transmitted, is exemplary ofthis invention. It is not intended that the scope of the invention belimited to the exemplary embodiments. For example, the digital datacould be transmitted alone, without linear summing with the analog FMsignal in various environments, thereby providing simple digital datatransmission from one point to another whether by wire or throughatmospheric free space. Such digital data transmission is useful insystems such as digital data storage and retrieval systems, wired andwireless LANS, microwave digital communication systems, cellulartelephone systems, etc. An exemplary embodiment would be thetransmission of a two-lobed digital signal as described herein in anenvironment having known colored interference at a particular frequency.The basis set may be spectrally shaped in such a case so that the twolobes surround the frequency at which the interference (e.g.,conventional microwave interference) is present. Additionally, the basisset of orthogonal waveforms output from generator 37 need not bespectrally shaped into two separate and spaced apart lobe or passbandportion; instead, the waveforms may be shaped into a single lobe portioncontaining substantially all of the digital data to be transmitted. Suchspectral shaping would be advantageous in environments not havinginterference such as the above mentioned analog FM signal of the firstembodiment or any other type of colored interference.

Additionally, dynamic interference cancelling may be utilized in certainembodiments so as to filter out specific portions of the received signalin environments having strong colored interference. Such interferencecancelling filters may be implemented following equalizer 113 oralternatively as part of equalizer 113. The taps of equalizer 113 may beprogrammed in such a manner that the equalizer itself filters out thecolored interference. Once given the above disclosure, many otherfeatures, modifications and improvements will become apparent to theskilled artisan. Such other features, modifications, and improvementsare, therefore, considered to be a part of this invention, the scope ofwhich is to be determined by the following claims.

What is claimed is:
 1. A method for preparing data for transmission froma first location to a second location through a medium, the methodcomprising:modulating the data with a basis signal set to generate aplurality of modulated signals, one modulated signal for each sequencein the basis signal set, wherein the basis signal set includesaplurality of sequences of a predetermined length, each of the sequencesdefining a waveform representing a digital data carrier, wherein each ofthe waveforms has predetermined desirable autocorrelationcharacteristics, and wherein each of the sequences making up the basissignal set is substantially orthogonal to other sequences within thebasis signal set.
 2. The method of claim 1 wherein each of the waveformshas an autocorrelation value approaching zero for substantially allpositive and negative time shifts, and a maximum autocorrelation valuefor no time shift.
 3. The method of claim 1 wherein acrosscorrelation/autocorrelation ratio of waveforms in the basis signalset is less than about 0.003.
 4. The method of claim 1 wherein acrosscorrelation/autocorrelation ratio of waveforms in the basis signalset is less than about 0.001.
 5. The method of claim 1 wherein each ofthe waveforms covers substantially an entire available frequency band.6. The method of claim 1 wherein each of the waveforms occupies a finitesignal bandwidth.
 7. The method of claim 1 wherein the basis signal setincludes at least 48 sequences.
 8. The method of claim 1 wherein each ofthe waveforms is spectrally shaped by a shaping function.
 9. The methodof claim 8 wherein the shaping function has a bandpass response.
 10. Themethod of claim 9 wherein the bandpass response has a null at a centerfrequency and passbands on both sides of the center frequency.
 11. Themethod of claim 10 wherein each of the passbands is located within afrequency band extending from about 100 KHz to about 250 KHz from thecenter frequency.
 12. The method of claim 1 wherein each of thesequences comprises multi-value samples, each multi-value sample havinga particular value selected from a set of three or more possible values.13. The method of claim 1 wherein the sequences are generated usingsingular value decomposition.
 14. A transmitter for transmitting datafrom a first location to a second location through a medium, thetransmitter comprising:a modulator configured to receive the data and abasis signal set, and to modulate the data with the basis signal set togenerate a plurality of modulated signals, one modulated signal for eachsequence in the basis signal set, wherein the basis signal set includesaplurality of sequences of a predetermined length, each of the sequencesdefining a waveform representing a digital data carrier, wherein each ofthe waveforms has predetermined desirable autocorrelationcharacteristics, and wherein each of the sequences is substantiallyorthogonal to other sequences within the basis signal set.
 15. Thetransmitter of claim 14 wherein each of the waveforms has anautocorrelation value approaching zero for substantially all positiveand negative time shifts, and a maximum autocorrelation value for notime shift.
 16. The transmitter of claim 14 wherein acrosscorrelation/autocorrelation ratio of waveforms in the basis signalset is less than about 0.003.
 17. The transmitter of claim 14 wherein acrosscorrelation/autocorrelation ratio of waveforms in the basis signalset is less than about 0.001.
 18. The transmitter of claim 14 whereineach of the waveforms covers substantially an entire available frequencyband.
 19. The transmitter of claim 14 wherein each of the waveformsoccupies a finite signal bandwidth.
 20. The transmitter of claim 14wherein the basis signal set includes at least 48 sequences.
 21. Thetransmitter of claim 14 wherein each of the waveforms is spectrallyshaped by a shaping function.
 22. The transmitter of claim 21 whereinthe shaping function has a bandpass response.
 23. The transmitter ofclaim 22 wherein the bandpass response has a null at a center frequencyand passbands on both sides of the center frequency.
 24. The transmitterof claim 23 wherein each of the passbands is located within a frequencyband extending from about 100 KHz to about 250 KHz from the centerfrequency.
 25. The transmitter of claim 14 wherein each of the sequencescomprises multi-value samples, each multi-value sample having aparticular value selected from a set of three or more possible values.26. The transmitter of claim 14 wherein the sequences are generatedusing singular value decomposition.